Double-ended forward converter and power supply device

ABSTRACT

A double-ended forward converter includes: a first switching element and a second switching element that are coupled to a primary side of a transformer; a pulse generation circuit that generates a pulse signal for controlling the first and second switching elements; an isolation transformer that converts the pulse signal into an alternating-current signal; a rectifier circuit that rectifies the alternating-current signal and generate gate voltages of the first and second switching elements; a driver circuit that includes a third switching element which drives gates of the first and second switching elements, a voltage generated on a secondary side of the isolation transformer being input to a gate of the third switching element; and a minus bias generation circuit that generates a source voltage of the third switching, based on a change in the voltage generated on the secondary side of the isolation transformer.

CROSS-REFERENCE TO RELATED APPLICATION

This application is a continuation application of InternationalApplication PCT/JP2013/001975 filed on Mar. 22, 2013 and designated theU.S., the entire contents of which are incorporated herein by reference.

FIELD

The embodiments discussed herein are related to a double-ended forwardconverter including a normally-on switching element.

BACKGROUND

In recent years, saving of energy resources in various fields hasattracted attention, and the influence thereof has spread to the fieldof, for example, power supply. Specifically, the higher efficiency of,for example, a switching power supply has been desired.

The switching power supply converts an input direct-current voltage to adesired direct-current voltage by using a direct current (DC)-DCconverter and outputs the desired direct-current voltage as a stabilizedpower supply voltage.

The following is a reference document.

-   [Document 1] Japanese Laid-open Patent Publication No. 2005-65393.

SUMMARY

According to an aspect of the invention, a double-ended forwardconverter includes: a first switching element and a second switchingelement that are coupled to a primary side of a transformer; a pulsegeneration circuit that generates a pulse signal for controlling thefirst and second switching elements; an isolation transformer thatconverts the pulse signal into an alternating-current signal; arectifier circuit that rectifies the alternating-current signal andgenerate gate voltages of the first and second switching elements; adriver circuit that includes a third switching element which drivesgates of the first and second switching elements, a voltage generated ona secondary side of the isolation transformer being input to a gate ofthe third switching element; and a minus bias generation circuit thatgenerates a source voltage of the third switching, based on a change inthe voltage generated on the secondary side of the isolationtransformer.

The object and advantages of the invention will be realized and attainedby means of the elements and combinations particularly pointed out inthe claims.

It is to be understood that both the foregoing general description andthe following detailed description are exemplary and explanatory and arenot restrictive of the invention, as claimed.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is a circuit diagram of a general double-ended forward converter;

FIGS. 2A and 2B are diagrams illustrating an example of a gate drivecircuit based on a transformer drive system;

FIG. 3 is a gate drive circuit diagram of an embodiment;

FIGS. 4A to 4D are diagrams for explaining an operation of a driverminus power supply circuit of an embodiment;

FIGS. 5A to 5G are simulation waveform diagrams of the gate drivecircuit of an embodiment;

FIG. 6 is a circuit diagram in which the gate drive circuit of anembodiment is applied to a double-ended forward converter;

FIGS. 7A to 7D are simulation waveform diagrams of a circuit in whichthe gate drive circuit of an embodiment is applied to the double-endedforward converter; and

FIG. 8 is a circuit diagram of a power supply device.

DESCRIPTION OF EMBODIMENTS

FIG. 1 illustrates a circuit diagram of a general double-ended forward(bipolar) converter serving as a type of DC-DC converter.

A double-ended forward converter 10 illustrated in FIG. 1 steps down aninput direct-current voltage VIN, thereby generating an outputdirect-current voltage Vout of a desired electric potential.

The input direct-current voltage VIN is a direct-current voltage of 385V obtained by converting an AC voltage of 80 (V) to 265 (V) to adirect-current voltage by using a switching power supply in, forexample, a server or the like. In addition, furthermore, in order to beused within the server, the input direct-current voltage VIN is steppeddown to a predetermined voltage by the double-ended forward converter10. For safety, based on a standard such as IEC60950, an AC input issupposed to be isolated from a direct-current voltage used within thedevice. Therefore, both the primary-side VIN generated from the AC inputand the secondary-side Vout use the double-ended forward converter 10based on a method of being isolated using a transformer T1.

The double-ended forward converter 10 includes primary-side switchesSW10 and SW12, feedback diodes D10 and D12, the transformer T1, aninductor L10, a smoothing capacitor C10, and synchronous rectifiers D14and D16.

For example, metal oxide semiconductor field-effect transistors(MOSFETs) are used for the primary-side switches SW10 and SW12. In thedouble-ended forward converter 10, if the primary-side switches SW10 andSW12 are simultaneously subjected to switching and caused to performon-off actions, thereby applying a current to the primary side of thetransformer T1, alternating-current power is generated on the secondaryside of the transformer T1. The alternating-current power is rectifiedby the first and second rectifiers D14 and D16 and is smoothed by thechoke coil L10 and the output smoothing capacitor C10, thereby beingconverted to the output direct-current voltage Vout. Note thatsynchronous rectification in which the first and second rectifiers D14and D16 are replaced with low-resistance FETs in order to reduce a lossis used in some cases.

Since the transformer T1 is excited in only one of two directions, acoil of the transformer T1 stores energy as soon as the primary-sideswitches SW10 and SW12 are turned off. Therefore, energy is fed back bythe feedback diodes D10 and D12, thereby resetting magnetic fluxes.

The on-off actions of the primary-side switches SW10 and SW12 arecontrolled by a pulse width modulation (PWM) signal. A circuit thatgenerates the PWM signal uses the output direct-current voltage Voutserving as a power supply on the secondary side of the transformer T1.Therefore, the PWM signal is converted by transformers T2 and T3 are isinput to gates of the switches SW10 and SW12 on the primary side of thetransformer T1. From this, the switches SW10 and SW12 on the primaryside are isolated from the PWM signal generation circuit on thesecondary side.

In order to reduce a power loss in the DC-DC converter, it is desirableto use a switching element whose on-resistance is low and whoseswitching speed is high. A high electron mobility transistor (HEMT) hasbeen developed, the HEMT using gallium nitride (GaN) serving as acompound semiconductor material that is not silicon. Hereinafter, thisswitching transistor will be called a GaN-HEMT.

Since, compared with silicon, having high electron mobility and highmutual conductance, many compound semiconductors each have a featurethat it is possible to reduce the on-resistance and capacitancegenerated between individual terminals of a transistor is low.

However, a silicon MOSFET of the related art is a normally-off type(enhancement type) in which turnoff is caused in a state of applying novoltage to a gate, whereas, usually the GaN-HEMT is, by contrast, anormally-on type (depression type) in which turns on in a state ofapplying no voltage to a gate. Therefore, in order to subject theGaN-HEMT to switching, a voltage is applied based on a negative powersupply circuit.

With reference to FIGS. 2A and 2B, a problem area in a case where theMOSFETs of the primary-side switches SW10 and SW12 are replaced with theGaN-HEMTs in the double-ended forward converter 10 will be described.

FIG. 2A illustrates an example of a gate drive circuit based on atransformer drive system, which controls a gate of a switching element.The gate drive circuit, based on a transformer drive system andillustrated in FIG. 2A, includes a PWM signal generation circuit 20, adamping resistor R3, a direct-current cut capacitor C3, an isolationtransformer T4, and a driver circuit. The damping resistor R3 limits acurrent so that the direct-current cut capacitor C3 and the isolationtransformer T4 do not resonate. The direct-current cut capacitor C3 cutsa direct current so that a current does not continue to flow owing tosaturation of the primary side of the isolation transformer T4. Theisolation transformer T4 isolates a PWM signal from the PWM signalgeneration circuit 20 and transmits the PWM signal to the secondary sidethereof. The PWM signal transmitted to a secondary-side air is input tothe gate of the switch SW10 and controls the gate of the switch SW10.

FIG. 2B illustrates waveforms of a gate voltage VG of the switchingelement. With reference to FIG. 2B, respective waveforms illustratechanges in the gate voltage VG of the switching element in a case wherethe duty ratio of the PWM signal is set to 10%, 50%, and 80%.

If the gate of the switching element SW10 is controlled by the circuit,based on a transformer drive system and illustrated in FIG. 2A, both apeak value and a zero level electric potential of the gate voltage VGfluctuate depending on the duty ratio of the PWM signal, as illustratedin FIG. 2B.

If, in this way, the gate voltage VG fluctuates depending on the dutyratio of the PWM signal, there is a possibility that it is difficult toperform a switching operation on the normally-on type GaN-HEMT.

Suitable embodiments according to the present disclosed technology willbe described in detail with reference to drawings.

FIG. 3 is a circuit diagram illustrates one embodiment of a gate drivecircuit that is based on a transformer drive system and that drives anormally-on type switching element.

In FIG. 3, a same symbol is assigned to a configuration element equal toor equivalent to the gate drive circuit, based on a transformer drivesystem and illustrated in FIG. 2A, and the description thereof will beomitted.

A gate drive circuit 70, based on a transformer drive system andillustrated in FIG. 3, includes the PWM signal generation circuit 20, anisolated transmission circuit 30, a half-wave voltage doubler rectifiercircuit 40, a driver circuit 50, and a driver minus power supply circuit60.

The PWM signal generation circuit 20 is equivalent to a PWM signalgeneration circuit used in a general step-down type converter andperforms switching on a normally-on type switching element SW30 by usingthe generated PWM signal.

The half-wave voltage doubler rectifier circuit 40 is a circuit forrectifying a secondary-side voltage Vtrans generated on the secondaryside of the gate transformer T4 and generates given voltages V+ and V−independent of the duty ratio of the PWM signal. The given voltages V+and V− are determined by the secondary-side voltage Vtrans of the gatetransformer T4 and a capacity ratio between two capacitors C2 and C4 asfollows.

V+=Vtrans×C2/(C2+C4)

V−=−Vtrans×C4/(C2+C4)

The driver circuit 50 is a circuit for supplying the gate voltage Voutto a gate of the switching element SW30 and outputs, to the gate of theswitching element SW30, one of the given voltages V+ and V− generated bythe above-mentioned half-wave voltage doubler rectifier circuit 40. AnNPN type transistor Q1 on the output side of the driver circuit 50 isturned on when a base voltage thereof becomes high, and the NPN typetransistor Q1 supplies the given voltage V+ on a collector side thereofto the gate of the switching element SW30. By setting the given voltageV+ to a voltage that exceeds a threshold value of the normally-on typeswitching element SW30, the switching element SW30 is put into anon-state.

In addition, a PNP transistor Q4 is turned on when a base voltagethereof becomes low, and the PNP transistor Q4 supplies the givenvoltage V− on an emitter side thereof to the gate of the switchingelement SW30. By setting the given voltage V− to a voltage less than orequal to a threshold value of the normally-on type switching elementSW30, the switching element SW30 is put into an off-state.

The base voltages of the NPN type transistor Q1 and the PNP transistorQ4 are controlled by two switching elements Q3 and Q5 configured in twostages.

The secondary-side voltage Vtrans of the gate transformer T4 is appliedto a gate of the first-stage switching element Q3.

The driver minus power supply circuit 60 generates a source voltage Vsof the switching element Q3, which varies depending on the duty ratio ofthe secondary-side voltage Vtrans.

Next, using FIGS. 4A to 4D, an operation of the driver minus powersupply circuit 60 will be described. FIG. 4A illustrates a circuit inwhich only a portion 90, which relates to generation of a driver minuspower supply and surrounded by a dashed line, is extracted from the gatedrive circuit 70, based on a transformer drive system and illustrated inFIG. 3.

Roles of individual components in this circuit are as follows.

R3: a damping resistor to limit a current so that the C3 and the T4 donot resonate.

C3: to cut a direct current so that a current does not continue to flowowing to saturation of the primary side of the T4.

T4: to isolate and transmit the PWM signal to the secondary side.

D3: rectifier diode.

C5: smoothing capacitor.

First, a reason why VS has a dependence property with respect to a dutyratio will be described. If being drawn by focusing on the primary sidein FIG. 4A, an equivalent circuit may be illustrated by an LC equivalentcircuit illustrated in FIG. 4B. This case is simplified under theassumption that since having a small value, the damping resistor R3 haslittle influence.

If a pulsed voltage having a peak voltage Vin is applied from a voltagesource V1, a current ILP1 ₁ flowing through LP1 ₁ monotonicallyincreases during a time period (D×T), during which a voltage is applied,and monotonically decreases in a state (D×T to T) of a zero voltage, asillustrated in FIG. 4C. So as to clarify a relationship with a dutyratio, currents in the respective time periods will be calculated.

A current change ΔI_(L(DT)) in the “D×T” time period is calculated asfollows.

$V_{L} = {{V_{in} - V_{S}} = {L_{P\; 1}\frac{{di}_{L}}{dt}}}$${L_{P\; 1}\frac{{di}_{L}}{dt}} = {\frac{\Delta \; i_{L}}{\Delta \; t} = {\frac{\Delta \; i_{L}}{DT} = \frac{V_{in} - V_{C}}{L_{P\; 1}}}}$${\Delta \; i_{L{({DT})}}} = {\frac{V_{in} - V_{C}}{L_{P\; 1}}{DT}}$

A current change ΔI_(L(DT to T)) in the “D×T to T” time period iscalculated as follows.

$V_{{LP}\; 1} = {{- V_{C}} = {L_{P\; 1}\frac{{di}_{{LP}\; 1}}{dt}}}$$\frac{{di}_{L}}{dt} = {\frac{\Delta \; i_{{LP}\; 1}}{( {1 - D} )T} = {- \frac{V_{C}}{L_{P\; 1}}}}$${\Delta \; i_{L{({{DT} \sim T})}}} = {{- \frac{V_{C}}{L_{P\; 1}}}( {1 - D} )T}$

Since, in a steady state, adding Δi _(L(DT)) to Δi_(L(DT to T)) becomeszero, Δi_(L(DT))+Δi_(L(DT to T))=0

In other words,

${{( \frac{V_{in} - V_{C}}{L_{P\; 1}} ){DT}} - {\frac{V_{in}}{L_{P\; 1}}( {1 - D} )T}} = 0$

If collecting V_(C),

V_(C)=V_(in)D

If a voltage of the voltage source V1, which temporally changes, isv_((t)), a voltage V_(LP1) between two end portions of L_(P1) isexpressed as follows.

V _(LP1) =v _((t)) −V _(C) =V _((t)) −V _(in) D

If being graphically illustrated, FIG. 4D is obtained, a peak voltage ofV_(LP1) is V_(in)−V_(in)D, and a voltage on a negative side thereof is−V_(in)D.

FIGS. 5A to 5G illustrate simulation waveform diagrams of the gate drivecircuit, based on a transformer drive system and illustrated in FIG. 3.

If, with reference to FIG. 5A and FIG. 5B, the duty ratio of theprimary-side voltage Vin of the gate transformer T4 increases, theprimary-side voltage Vin being amplitude-modulated by the PWM signal,the peak value of the secondary-side voltage Vtrans decreases. Inaddition, depending on the duty ratio of the primary-side voltage Vin,the zero voltage electric potential of the secondary-side voltage Vtransvaries.

With reference to FIG. 5C and FIG. 5D, FIG. 5C and FIG. 5D illustratethe given voltages V+ and V−, respectively, generated by the half-wavevoltage doubler rectifier circuit 40 by rectifying the secondary-sidevoltage Vtrans of the gate transformer T4. It may be confirmed that thegiven voltages V+ and V− are determined by the secondary-side voltageVtrans of the gate transformer T4 and the capacity ratio between the twocapacitors C2 and C4 and are not dependent on the duty ratio of the PWMsignal.

In a case where, with reference to FIG. 5E and FIG. 5F, the duty ratioof the PWM signal is 10%, a zero level of the gate voltage VG of thefirst-stage switching element Q3 is 31 1.2 V, whereas, by contrast, thesource voltage VS=−12 V×0.1=−1.2 V is satisfied. Therefore, a voltagedifference between the gate and the source becomes zero. In a case wherethe duty ratio of the PWM signal is 50%, the zero level of the gatevoltage VG of the first-stage switching element Q3 is −6 V, whereas, bycontrast, contrast the source voltage VS=−12 V×0.5=−6 V is satisfied.Therefore, the voltage difference between the gate and the sourcebecomes zero. In a case where the duty ratio of the PWM signal is 80%,the zero level of the gate voltage VG of the first-stage switchingelement Q3 is −9.6 V, whereas, by contrast, the source voltage VS=12V×0.8=−9.6 V is satisfied. Therefore, the voltage difference between thegate and the source becomes zero.

In this way, even if the duty ratio of the PWM signal changes, therebycausing the zero level of the gate voltage VG of the switching elementQ3 to change, it becomes possible to continuously set the voltagedifference between the gate and the source to a given level by changingthe source voltage VS in accordance with a change in the zero level ofthe gate voltage VG.

According to the present embodiment, a given voltage, generated by thehalf-wave voltage doubler rectifier circuit 40 and independent of theduty ratio of the PWM signal, is supplied to the gate of the normally-onGaN-HEMT on the primary side of a transformer in the double-endedforward converter. Therefore, the on-off action is correctly performed.In addition, even if the duty ratio of the PWM signal changes, it ispossible to set, to a given level, a gate-to-source voltage V of theswitching element Q3 in the drive circuit that controls the normally-onGaN-HEMT. Therefore, it is possible to perform a stable action.

FIG. 6 illustrates a circuit diagram of a DC-DC converter in which eachof two switches included in a double-ended forward circuit 80 is drivenby the gate drive circuit 70, based on a transformer drive system andillustrated in FIG. 3.

FIGS. 7A to 7D illustrate simulation waveforms of main portions of theDC-DC converter illustrated in FIG. 6. FIG. 7A illustrates a gatevoltage VG of a switching element SW10, FIG. 7B illustrates a draincurrent IDS, FIG. 7C illustrates an inductor current IL of an inductoron an output side, and FIG. 7D illustrates a simulation waveform of a DCoutput voltage Vout. With reference to FIG. 7A and FIG. 7B, it isconfirmed that the gate voltage VG of the normally-on GaN-HEMT falls toan electric potential of about −2 V or less at the time of being low andthe flow of the drain current IDS is stopped, thereby correctlyperforming the on-off action.

Note that additionally the gate drive circuit, based on a transformerdrive system, of an embodiment is able to be applied to a double-endedflyback type DC-DC converter.

FIG. 8 illustrates an example of a circuit diagram of a switching powersupply device in a server or the like.

The power supply device illustrated in FIG. 8 includes a rectifiercircuit 110, a PFC circuit 120, a control unit 150, and a DC-DCconverter 160.

The rectifier circuit 110 is a diode bridge in which four diodes areconnected in a state of bridge. The rectifier circuit 110 is connectedto an alternating-current power source 100 and full-wave-rectifies andoutputs alternating-current power. Here, since the output voltage of thealternating-current power source 100 is Vin, the input voltage of therectifier circuit 110 is Vin. The rectifier circuit 110 outputs electricpower obtained by full-wave-rectifying the alternating-current powerinput from the alternating-current power source 100.

Since the alternating-current power whose voltage is, for example, 80(V) to 265 (V) is input to the rectifier circuit 110, voltage drops inthe four diodes in the rectifier circuit 110 are negligible. Therefore,the output voltage of the rectifier circuit 110 is regarded as Vin.

The PFC circuit 120 includes an inductor, a switching element, a diode,and a smoothing capacitor, which are connected in a T shape, and is anactive filter circuit that reduces a distortion such as a harmonic waveincluded in a current rectified by the rectifier circuit 110 and thatimproves a power factor of electric power.

For example, a boosting inductor is used for the inductor, and forexample, a metal oxide semiconductor field-effect transistor (MOSFET) isused for the switching element. The control unit 150 applies a pulsedgate voltage to a gate of the switching element, thereby performing theon-off action of the switching element, and the switching element ispulse-width-modulation (PWM)-driven.

The diode only has to have a rectification direction oriented in adirection from the inductor to the smoothing capacitor and, for example,a fast recovery diode or a SiC schottky diode is used therefor.

The control unit 150 outputs the pulsed gate voltage to be applied tothe gate of the switching element. Based on the voltage value Vin of thefull-wave-rectified electric power output from the rectifier circuit110, a current value IQ of a current flowing through a switching element22, and a voltage value Vout on the output side of the smoothingcapacitor, the control unit 150 determines the duty ratio of the gatevoltage and applies the gate voltage to the gate of the switchingelement. As the control unit 150, a multiplier circuit capable ofcalculating the duty ratio, based on, for example, the current value IQand the voltage values Vout and Vin, may be used.

The smoothing capacitor smooths a voltage output by the PFC circuit 120and inputs the smoothed voltage to the DC-DC converter 60.

The double-ended forward type converter of the present embodiment,described above, is used for the DC-DC converter 160.

Direct-current power whose voltage is, for example, 385 (V) is input tothe DC-DC converter 160, is converted by the DC-DC converter 160 intodirect-current power of, for example, 12 (V), and is output to a loadcircuit 170.

All examples and conditional language recited herein are intended forpedagogical purposes to aid the reader in understanding the inventionand the concepts contributed by the inventor to furthering the art, andare to be construed as being without limitation to such specificallyrecited examples and conditions, nor does the organization of suchexamples in the specification relate to a showing of the superiority andinferiority of the invention. Although the embodiments of the presentinvention have been described in detail, it should be understood thatthe various changes, substitutions, and alterations could be made heretowithout departing from the spirit and scope of the invention.

What is claimed is:
 1. A double-ended forward converter comprising: afirst switching element and a second switching element that are coupledto a primary side of a transformer; a pulse generation circuit thatgenerates a pulse signal for controlling the first and second switchingelements; an isolation transformer that converts the pulse signal intoan alternating-current signal; a rectifier circuit that rectifies thealternating-current signal and generate gate voltages of the first andsecond switching elements; a driver circuit that includes a thirdswitching element which drives gates of the first and second switchingelements, a voltage generated on a secondary side of the isolationtransformer being input to a gate of the third switching element; and aminus bias generation circuit that generates a source voltage of thethird switching, based on a change in the voltage generated on thesecondary side of the isolation transformer.
 2. The double-ended forwardconverter according to claim 1, wherein the first and second switchingelements are GaN-HEMTs.
 3. The double-ended forward converter accordingto claim 1, wherein the rectifier circuit is a half-wave voltage doublerrectifier circuit.
 4. The double-ended forward converter according toclaim 1, wherein the pulse signal for controlling the first and secondswitching elements is a PWM signal, and a duty ratio varies based on anoutput of the double-ended forward converter.
 5. A power supply devicecomprising: an alternating-current power source; a rectifier circuitthat rectifies a current of the alternating-current power source; apower factor improvement circuit that smooths the current rectified bythe rectifier circuit and generate a first direct-current voltage; and aDC-DC converter that generates a second direct-current voltage from thefirst direct-current voltage, wherein the DC-DC converter includes firstand second switching elements that are coupled to a primary side of atransformer, a pulse generation circuit that generates a pulse signalfor controlling the first and second switching elements, an isolationtransformer that converts the pulse signal into an alternating-currentsignal, a rectifier circuit that rectifies the alternating-currentsignal and generate gate voltages of the first and second switchingelements, a driver circuit that includes a third switching element whichdrives gates of the first and second switching elements, a voltagegenerated on a secondary side of the isolation transformer being inputto a gate of the third switching element, and a minus bias generationcircuit that generates a source voltage of the third switching, based ona change in the voltage generated on the secondary side of the isolationtransformer.